Method and apparatus for digital detection of electronic markers using frequency adaptation

ABSTRACT

An electronic marker locator with a digital architecture for providing accurate and consistent estimation of the signal strength is presented. The marker locator includes a Digital Phase-Locked Loop (DPLL) structure. The electronic marker locator transmits known and adjustable frequency bursts corresponding to the markers to be located while synchronously capturing the signals returned from the markers. Because of the convergence properties of the DPLL, very consistent measurements of the reflected marker signal field strength are possible, resulting in both an improvement of maximum detection depth and depth accuracy. Further, the analog front-end hardware can be reduced, offering wider resistance to component tolerances, lower calibration and test times, and flexible frequency selectivity.

BACKGROUND

1. Field of the Invention

The present invention is directed toward detection of concealedelectronic markers that are commonly buried alongside pipes or cablesand, in particular, to a digital implementation of a combined pipe andcable locator device to simultaneously search for buried markers withhigh accuracy and repeatability.

2. Discussion of Related Art

Utility conduits are often buried underground or concealed in walls andtherefore are not readily accessible or identifiable. It is oftennecessary to locate these concealed utility conduits in order to repairand replace them. It is also important to know the location of utilityconduits in order to avoid them while excavating an area. Examples ofhidden utility conduits include pipelines for gas, sewage, or water andcables for telephone, television, or power.

There are various ways to locate concealed objects, for example, usingline locators or marker locators. Line locators are appropriate whenseeking electrically conductive objects, such as metallic pipelines andcables. Line locators may also be used for finding non-electricallyconducting conduits when the conduit is marked with a conducting tracewire or trace tape buried along the conduit. The process of applying anAC signal to the conductor at an accessible point and detecting theresulting electromagnetic radiation is well known in the art. When an ACsignal is applied, the conductor acts as an antenna radiating anelectromagnetic field along its entire length that can be detected by aline locator.

In such an application, a line locator used above ground detectselectromagnetic emissions from conductors underground. A disadvantagewith relying solely on the line locator device is that it may fail toidentify and distinguish among various types of utility conduits andconductors. Additionally, line locator devices cannot be used to locatenon-conductive lines, such as, for example, gas lines, fiber optic linesand plastic water lines when those non-conductive lines are not markedwith trace wires.

Conduits may also be marked with electronic markers, either at surfacelevel or underground. Buried electronic markers may be used to locateand identify a number of concealed objects such as cables, pipes, accesspoints, underground stock piles, survey points and septic tanks.Typically, marker locators locating passive, active, or smart markersgenerate an electromagnetic field that induces a response in the markerthat can be monitored by a detector of the marker locator. Again,significant difficulty in marker type identification and depthdetermination may result, especially if multiple markers of differingtypes and depths are present.

A recent development is disclosed in U.S. Pat. No. 6,617,856 B1, whichdescribes a DSP based marker locator that substantially reducesdetection inaccuracies attributed to analog mixers, antenna saturation,and DC offsets common in many designs. A quadrature mixer and IIRadaptive filter are used to modify the integration (averaging) time ofthe system to enhance performance when the markers are deeply buried,and to allow a more responsive mode (shorter averaging times) when thisis not the case.

It is desirable in such marker detection systems that the results berepeatable and accurate. Therefore, methods of detecting the maximumreflected signal strength of a marker at a certain depth are important.It is also important that marker locators determine with a high degreeof accuracy the particular type of marker that is present. Therefore,there is a need to develop more highly accurate and reliable systems fordetecting marker types and marker depths.

SUMMARY OF THE INVENTION

In accordance with the present invention, a marker locator receiverdigital architecture is described that provides for digital marker andline location. In this fashion, highly accurate identification of markertype and depth determination can be accomplished.

A marker locator according to some embodiments of the invention includesat least one transmitter channel coupled to an electromagnetic fieldgenerator, at least one receiver channel coupled to an electromagneticfield detector, and a digital processor coupled to provide signals tothe at least one transmitter channel and to receive signals from the atleast one receiver channel. In the marker locator, the digital processorreceives signals from the at least one transmitter channel betweenapplication of signals to the at least one transmitter channel, thedigital processing system averaging signals received from the at leastone receiver channel to determine signal strength and the frequency ofsignals received.

A method of locating one or more markers according to some embodimentsof the invention include generating a series of electromagnetic pulses,receiving signals as a function of time between application of thepulses, averaging the signals over a predetermined number of pulses toobtain an average decay signal, initially determining a frequency, fieldstrength, and phase for responses from the one or more markers,determining the frequency, field strength, and phase more accurately,and refining the electromagnetic pulses in order to provide resonantfrequencies for each of the pulses.

Additionally, in some embodiments a dual-mode locator thatsimultaneously offers both line and marker location methods ispresented. Using such embodiments, an operator using conventional linetracing methods (with a separate transmitter to directly or inductivelycouple the signal at an access point) can also be warned of the presenceof one or more electronic markers in the same vicinity. Furthermore, insome embodiments multiple marker types can be searched forsimultaneously, and the user can be alerted to the presence of eachmarker type as encountered. When ambiguities in marker type and signalstrength exist due to a “neighbor detection” or “near-far” problem, itis advantageous for a marker locator to be able to reliably discriminatetargeted marker types from others using a quantitative method. Duringthe search phase of marker location (for example, a background markersearch activity while in line-locate mode), a frequency adaptationmechanism can be included to create the highest possible signal-to-noiseratio (SNR) in the reflected marker signal response. Finally in the lockphase, when an individual marker is known and locked onto by the markerlocator, accurate and repeatable signal strength estimates can beattained prior to invoking a depth calculation.

These and other embodiments are further discussed below with referenceto the following figures.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 illustrates a marker locator in operation by a locationtechnician.

FIG. 2A shows an example of a ball-type passive marker.

FIG. 2B shows an example of a disk-type passive marker.

FIG. 3 shows the electrical schematic diagram for a single LC circuit ofa passive marker.

FIG. 4 shows a block diagram of an embodiment of a combined line andmarker locator according to the present invention.

FIG. 5 illustrates an example of the relative phase differences in asuperposition of transmit signals from the perspective of a receiver ADCinput.

FIG. 6 shows a block diagram of an embodiment of a time averagecalculation used to estimate the marker response signal according to thepresent invention

FIG. 7 shows a block diagram of an embodiment of a frequency domainoperation that can be invoked on the averaged decay response todetermine which marker types are likely active in a current search.

FIG. 8 shows a block diagram of an embodiment of a successive markercancellation structure whereby detected marker types are sequentiallyremoved from the averaged decay signal.

FIG. 9 shows a block diagram of an embodiment of asingle-degree-of-freedom (SDOF) curvefit operation that can be invokedin a locked condition to create a modified decay for the next stage.

FIG. 10 shows a block diagram of an embodiment of a digital phase lockedloop (DPLL) that can be used to update the marker frequency according tothe present invention.

FIG. 11 shows a block diagram of an embodiment of a first ordernumerically controlled oscillator (NCO) of a DPLL such as that shown inFIG. 10, according to the present invention.

FIG. 12 shows a block diagram of an embodiment of a synthesis operationthat takes as input the result of the SDOF curvefit shown in FIG. 9 andcreates a synthesized time decay at the marker resonant frequency withstarting phase zero.

FIG. 13 shows a block diagram of a time reversal of the input signal,which can be a superposition of all active marker frequencies.

FIG. 14A, Case 1, illustrates a marker transfer function for scenariosin which only one active marker is present.

FIG. 14B, Case 2, illustrates a scenario in which three types of markersare present, and further where all types have approximately the sameamplitude.

FIG. 14C, Case 3, illustrates an example of the same physical scenarioas Case 2, but after a power control algorithm has increased theamplitude of marker type 2 relative to the other two types.

FIG. 15 illustrates a pulse stream and response in an embodiment of amarker locator system according to the present invention that detects asmart marker.

In the figures, elements having the same designation have the same orsimilar functions. Elements in the figures are not drawn to scale.

DESCRIPTION OF THE EMBODIMENTS

Generally, electronic markers consist of two types, namely, activemarkers and passive markers. Active markers radiate a signal detectableat the surface; however, they require a power source. Passive markers,on the other hand, require no power source and become active wheninduced by an external electromagnetic field, which can be generatedwith a portable power source.

A marker locator is a device for detecting and determining the locationof concealed or buried markers. Passive markers typically include amulti-turn wire loop (coil) tuned with a capacitor to a pre-determinedresonant frequency. A flexible implementation of an electromagneticmarker locator is described in U.S. patent application Ser. No.10/227,149, “Procedure and Device for Determining the Location of BuriedElectronic Markers,” by Hubert Schlapp and Richard Allin, which isherein incorporated by reference in its entirety. A fully digitalimplementation of an electromagnetic line locator is described in U.S.patent application Ser. No. 10/622,376, “Method and Apparatus forDigital Detection of Electromagnetic Signal Strength and SignalDirection in Metallic Pipes and Cables”, by James W. Waite and Johan D.Överby, which is herein incorporated by reference in its entirety.

FIG. 1 illustrates a marker locator according to the present inventionas operated by a location technician 6. Location technician 6 holdsmarker locator 1 directed towards ground level 7 to find the location ofhidden passive markers 10 and 12. The hidden passive markers 10 and 12can each be coded with a resonant frequency in order to identify thetype of utility lines 11 and 13 that each frequency respectively marks.

Commonly, a passive marker is the combination of a wire coil and acapacitor enclosed within a non-metallic protective enclosure. Thecombination creates an inductance-capacitance (LC) circuit defined by aninductance developed by the wire coil and a capacitance held by thecapacitor. The LC circuit operates in a resonance mode at its resonantfrequency f given by the equation: $\begin{matrix}{f = \frac{1}{2\pi\sqrt{LC}}} & \left( {{Equation}\quad 1} \right)\end{matrix}$where L is the inductance of the wire coil and C is the capacitance ofthe capacitor.

FIG. 2A shows an example of a ball-type passive marker. Passive marker10 is a spherical passive marker housing three LC circuits 10A, 10B, and10C. The coils of each LC circuit 10A, 10B, and 10C are positioned inorthogonal Cartesian planes such that the three tuned circuits produce auniform radio frequency (RF) field.

FIG. 2B shows a disk-type passive marker. Passive marker 12 is a flatpassive marker housing a single LC circuit 12A with the coil positionedin the horizontal X-Y plane.

FIG. 3 shows the electrical schematic diagram of a single LC circuit.The coil acts as an inductor 16, and is connected in parallel with acapacitor 18 to form a resonant tank circuit 14. The frequency f of thepassive marker is set by the resonant frequency of the passive LCcircuit, which can be tuned to a preset value.

Different types of utility lines are each associated with uniqueresonate frequency values. Markers with different resonant frequenciesmay also be colored for quick identification when installed. Sixdistinct frequency/color combinations are commonly used: 77.0 kHz(Orange/Black) for Canadian telephone and Cable TV; 83.0 kHz (Yellow)for Gas; 101.4 kHz (Orange) for Telephone; 121.6 kHz (Green) forSanitary/Waste water; 145.7 kHz (Blue) for Water; and 169.8 kHz (Red)for Power. Of course, these frequencies (and colors) have beendesignated by conventional use and are not meant to be restrictive.

Though passive electronic markers have several advantages over tracingwires, they are still subject to detection ambiguities. U.S. applicationSer. No. 10/227,149 (Schlapp, Allin) discloses methods of scanning formultiple marker types, with the goal to reduce the time consumed byseparate searches for each type of marker, or to provide alerts for thepresence of non-targeted markers. Additional search techniques arepresented with the aim of mitigating the “neighbor detection” problemwhere emissions of marker-types not being searched for overwhelm thereceiver producing false-positive indications, and the “near-far”problem where emissions from nearby markers can override signals fromthe farther placed marker possibly producing an erroneous markerindication. U.S. application Ser. No. 10/227,149 further disclosesmethods of sequentially testing for the presence of adjacent markers andusing the measured signal levels to deduce whether or not targetedmarkers may be obscured by the “neighbor detection” or “near-far”problems.

U.S. patent application Ser. No. 10/227,149 discloses the possibility ofinvoking a parallel search method in which multiple marker types can beexcited by a single transmitter pulse. Specific detection methods arediscussed below that allow discrimination of specific marker types fromthe collective marker response to such a transmit signal.

Optimized accuracy of measuring both the marker type and depth can beachievable where the marker LC circuit is repeatedly stimulated at afrequency that is precisely matched to the natural frequency of themarker. In Equation 1, the marker natural frequency f is subject tomanufacturing tolerances in the inductive and capacitive elements, thuscan vary from marker to marker (of the same type) by a few kHz. A markersignal strength detection device, then, should include some adaptationor search mechanism for finding the natural resonant frequency of amarker in order to increase accuracy of measurement. Once the naturalfrequency is determined for a specific electronic marker, a consistentmeasure of the signal strength is attainable at the best possiblesignal-to-noise ratio (SNR).

A block diagram of a combined marker and line locator 400 according tosome embodiments of the present invention is shown in FIG. 4. Locator400 includes a line locator section 424 and a marker locator section423. In the embodiment shown in FIG. 4, the top two data paths thatoriginate from the reference antenna 401 and top antenna 402 areassociated with a simple peak-mode line locator 424. The signal detectedby reference antenna 401 can be processed through an amplifier 441 and afilter 442 before being digitized by analog-to-digital converter (ADC)443. Similarly, the signal detected by antenna 402 can be processedthrough amplifier 444 and filter 445 before being digitized by ADC 446.The digital signals from ADC 443 and ADC 446 are input to a processor420. The signal from ADC 443 is received by channel processor 447 andthe signal from ADC 445 is received by channel processor 448. Channelprocessors 447 and 448 determine signal strengths and signal directions,which can be displayed on locator display 450 through interface 449. Insome embodiments, processor 420 may include a microprocessor ormicrocontroller executing software for performing the functions ofprocessor channels 447 and 448. Measurements of signal strength fromantennas 401 and 402, which are typically separated vertically by aknown amount, allow calculation in processor 420 of conductor depth andthe current through the conductor. In accordance with the presentinvention, any line locator structure may be included in line locator424. For example, a line locator with left/right indication is disclosedin U.S. Pat. No. 6,407,550, issued on Jun. 18, 2002 to Parakulam et al.and assigned to the same assignee as is the present disclosure, which isherein incorporated by reference in its entirety.

Marker locator section 423 according to some embodiments of the presentinvention includes an antenna 406 coupled to receive a transmitteroutput signal 422 and a receiver input signal 421. Marker locatorsection 423 is capable of transmitting electromagnetic radiation throughmarker antenna 406 at one or more of a plurality of fixed frequencies.Further, receiver input signal 421 can include signals detected bymarker antenna 406 at one or more of a plurality of fixed frequencies.Marker locator 423 further includes a processor 420 coupled to receivethe receiver input signal 421 for analysis and to generate thetransmitter output signal 422.

In some embodiments, processor 420 can include a fixed-point digitalsignal processor (DSP). In such a DSP, most if not all marker specificsignal generation and detection algorithms can be implemented insoftware. Further, some or all of the functions of line locator section424 can be performed by the DSP of processor 420. Receiver input signal421 is input to switch 425. When switch 425 is engaged to recognizereceiver input signal 421, receiver input signal 421 is then amplifiedin amplifier 426, filtered in filter 427, and digitized in ADC 428before being input to processor 420. Additionally, transmitter outputsignal 422, digitally generated by processor 420, is input todigital-to-analog converter 403, filtered in filter 404, and amplifiedin amplifier or driver 405 before being received by switch 407. Whenswitch 407 is engaged, transmitter output signal 422 is applied tomarker antenna 406. In some embodiments, filters 427 and 404 can both below-pass filters. Filter 427 then acts as an anti-aliasing filter whilefilter 404 acts as a reconstruction filter.

As shown in FIG. 4, the only hardware in marker locator section 423 arethe analog filters, amplifiers, and switches that control signalsarriving at or coming from marker antenna 406. In addition, line locatorreceiver section 424 also includes amplifiers and filters. Thus, markerlocator section 423 according to the present invention results insignificantly reduced analog front-end hardware requirements, a wideresistance to component tolerances, lower calibration and test time, andflexible frequency selectivity. Marker locators according to embodimentsof the present invention provide accurate and consistent estimation ofthe electronic marker signal strength in extremely noisy environments.

As is shown in FIG. 4, processor 420 receives receiver input signal 421into a time average block 430. Time average block 430 synchronouslyaverages the signals received at periodic intervals upon repeatedapplication of the transmitter output signal 422 to marker antenna 406.As such, time average 430 can measure the time delay of the returnedpulse with lower noise than without the average function. The outputsignal from time average 430 is input to marker search block 431. Markersearch block 431 uses frequency domain methods to extract signals in thetime average signal 430 that correspond to particular marker types. Whenlocked, phase calibration 432 and detection DPLL 433 have identified theresonant frequency of a marker and can output the resonant frequency andthe return signal strength for display on locator 450. Additionally, thefrequency generated either by phase calibration 432 or by detection DPLL433, selected in switch 434, is input to marker transmitter outputgenerator 435, which generates digitally the transmitter output signal422. Therefore, processor 420 locks marker locator 423 onto one or morefrequencies corresponding to the resonant frequencies of one or moremarkers that are being simultaneously detected in receiver input signal421.

Much of the performance improvement achieved in some embodiments of thepresent invention is primarily due to the frequency adaptation andselectivity resulting from a set of digital phase locked loops (DPLLs)implemented in detection DPLL 433. In some embodiments, detection DPLL433 can include one DPLL for each marker type being simultaneouslytracked. The tracking algorithm supports a successive detection processwherein markers that are buried deeper and whose signal may be obscuredby shallower markers of a different type are exposed throughcancellation of the stronger signals of the identified markers.

Each digital phase-locked loop in detection DPLL 433 can be a firstorder phase-locked loop that adapts only the frequency of a numericallycontrolled oscillator (NCO). There is no phase adaptation since this isknown and deterministic, based on the repeated output signal from asynchronous transmitter pulse of transmitter output signal 422. In someembodiments, the DPLL frequency update is performed only once peraveraged time decay from block 430 (corresponding to the time betweensuccessive pulses of transmitter output signal 422). The procedure ofupdating the frequency only once per pulse period of transmitter outputsignal 422 represents a significant reduction in the amount of data tobe processed through the DPLLs. In some embodiments, data processing inprocessor 420 may be reduced by a factor of between 10 and 500(depending on the pulse repetition rate). Therefore, in some embodimentsthe number of DSP instruction cycles utilized in the processor 420 ineach cycle can be minimized. Each DPLL frequency update is representedin a new output frequency block that is combined together bysuperposition in block 435.

In some embodiments, when multiple marker types are detected in markersearch 431 a power control algorithm can be implemented to enhance theprocess of successive cancellation of stronger marker signals and thushelp mitigate the near-far problem. The near-far problem occurs whensignals from marker types that are adjacent in frequency to a targetedmarker appear as if they originated from the targeted markers. The mostcommon example is when a marker closest to the transmitter (shallowestin the ground) overpowers a targeted marker because their respectiveresonant frequencies are similar. Because signal power dropsexponentially with distance and the marker frequency response isrelatively broad (as per the first order LC filter characteristic of thetypical marker), the wrongly-identified nearer marker can completelyhide the targeted farther marker. Embodiments of the present inventionallow adjustment of the transmitted signal power between the targetedmarker type and the adjacent markers in order to support removal of theenergy signal originating from the adjacent marker by successivecancellation of that energy from the received signal.

Another reason to implement power control is battery life—if thetransmitter were to continuously transmit at a power higher than thatneeded to maintain an acceptable SNR, the battery lifetime can begreatly reduced. Using a combination of power control and modificationof the transmitted signal burst rate, the marker locator may transmitusing the minimum power needed for maintaining the required SNR ratio,thus conserving its battery life.

In some embodiments, the reflected signal power levels of each markerreaching the receiver undergo constant changes because of persistentoperator movement of the marker locator with respect to the ground. Thesignal processing problem that the changing power levels represents iscompounded by the growing use of multiple marker types to tag utilitieswithin the same physical location (leading to instances of the near-farproblem). To overcome these obstacles, real-time tracking of multiplemarker types using digital phase locked loops, as well as power control,can be utilized. In some embodiments of the invention, as described inthe specification, both the frequency and amplitude of the marker searchsignals can be adapted to optimize the detection of multiple markertypes while avoiding many of the neighbor detection ambiguitiesprevalent in prior art marker locators.

In some embodiments of the present invention, a consistent signalstrength measurement is maintained to provide the maximum possible SNR.This arrangement can be achieved because the DPLL locks to the naturalresonant frequency of the marker. An increased confidence in theindications of depth and position provided by the marker location systemcan therefore be achieved.

Marker locator transceivers, such as marker locator section 423 oflocator 400, transmit known and adjustable frequency bursts whilesynchronously capturing the signals reflected from the markers. In theembodiment illustrated in FIG. 4, the output signal bursts are createdin transmitter output signal generator 435. For a simple searchinvolving only one type of marker, the initial output signal passed toDAC 403 is a single sinusoidal tone at the nominal marker frequency(e.g., 101.4 KHz for telephone cable markers). DAC 403 is also capableof transmitting a marker excitation signal combined by superpositionthat stimulates multiple markers simultaneously. In some embodiments, animportant constraint is placed on the transmitter output signal—thephase of the output burst is set such that the beginning of the inputblock after ADC sampling in the receiver is defined as zero degrees foreach marker stimulus frequency. Therefore, the digital phase locked loop(DPLL) in the receiver path (i.e., in detection DPLL 433) can create anerror signal and tend toward a locked state at zero-phase.

To ensure that correct phase adjustments are made to the transmittedsignal, the entire delay chain between DAC 403 and ADC 428 can becharacterized by a calibration operation 432. This synchronous stimulusresponse measurement procedure can be invoked when no markers arepresent in the environment. The result of the phase calibrationoperation 432 is that the appropriate group delay can be accounted forin the transmit burst. The tail end of a transmitter burst for twomarker frequencies is shown, for example, in FIG. 5, from theperspective of the data sampled at the ADC 428. The phase between thetwo signals is zero at time zero, corresponding to the first ADC sampleof the new burst. Not shown in FIG. 5 is the marker response data thatis collected starting at time zero.

Switches 407 and 425 (FIG. 4) limit the “self echo” signal seen in thereceive chain that is due to the transmitter directly, rather thanindirectly through the marker(s). These switches toggle between groundand signal depending on the phase of the pulse repetition clock 435,with the result that the receiver chain sees ground during the highlevel output of the transmitter. An instant after time zero (zero phaseat the ADC input), the transmitter output is grounded (through animpedance matching resistor), truncating the decaying impulse responseof filter 404 and keeping the transmit output signal 422 from impactingthe receiver measurement of the marker decay in receiver input signal421. Even with the switches in place some energy does leak from thetransmitter to the receiver, but this leakage energy can be within thelimits of the high gain analog circuitry of amplifier 426 and filter 427to sustain without saturation. The residual self-echo signal isdetermined by calibration operation 423 and later subtracted from theaveraged received marker decay signal.

Some embodiments of the invention allow the simultaneous excitation ofmultiple marker types, therefore transmitter signal generation 435 usesthe calibrated group delay values for each nominal marker centerfrequency from which to form the output signal superposition. Table 1below shows an example of the group delay corresponding to a non-linearphase characteristic for one embodiment of the analog anti-alias filter427. Because of phase dispersion, in FIG. 5 at time zero, the two zerophase marker stimulus signals input to ADC 428 will not have this samephase relationship at the signals output from transmitter DAC 403. Table1 represents the delays where filter 427 is an 8^(th) order Butterworthfilter. TABLE 1 Marker Type Frequency (kHz) Magnitude (dB) Group Delay(us) Cable TV 77.0 0 4.87 Gas 83.0 0 4.93 Telephone 101.4 0 5.19Sanitary 121.6 −0.01 5.65 Water 145.7 −0.15 6.73 Power 169.8 −1.41 8.16

During transmitter output generation 435, the group delays can be takeninto account for each marker stimulus signal and transmitter outputsignal 422 can be adjusted accordingly. The easiest way to performadjustments to transmitter output signal 422 is to generate the signalsin reverse time order (starting with the required phase per marker type)at time zero in the receiver. After the entire block is generated forall frequencies, it is fed to the DAC 403 backwards in time from theorder of creation. Thus the phase zero reference point is almost thelast sent out to DAC 403. In some embodiments, a small number of extrasamples can be created after time zero. Typically, the extra samples areapproximately equal to the group delay through the signal chain for eachmarker frequency.

Other embodiments of this invention can utilize higher sample rate ADCsfor ADC 428 that allow anti-alias filter implementations that areessentially linear phase in the range of marker frequencies. Yet otherembodiments can make use of a delta-sigma ADC 428 and DAC 403 thateliminate the need for external non-linear phase filters given theoversampling inherent in the delta-sigma approach. In any of these otherembodiments, the phase calibration task is simplified, but other factorsmay steer the designer away from them. For example, higher rate ADCsgenerally require more power, which limits their practicalimplementation in a battery operated marker locator. Furthermore thereis a tradeoff in the number of bits of precision of an ADC with themaximum sample rate. Even so, rapid technology improvements in thecommunications sector (especially digital subscriber line (DSL)technologies) are driving the development of integrated DAC and ADCdevices (codecs) that have compatible bandwidths with that of typicalelectronic marker detection systems. The DSL hybrid interface to twistedcopper pair transmission media, for example, requires a self-echocanceller and can provide a solution to the similar problem mentionedabove resulting from the marker transmit/receive antenna 406.

After calibration, in some embodiments using a time division between thetransmit pulse and marker response, each and every transmit burst hassubstantially the same zero phase (independent of marker type or groupdelay) when referenced at time zero (at the first ADC sample at ADC428). This key fact can be paired with another regarding the LC circuitsthat represent the markers: As the excitation frequency increases andcrosses the resonance point, the phase response of the marker goesthrough a 180° positive to negative shift. The phase is substantiallyzero at resonance. If an excitation frequency composed according to thedescription above also happens to be exactly equal to the naturalfrequency of the marker, the phase of the signal at sample zero from ADC428 will also be zero. Conversely if the excitation frequency issomewhat different from the marker's natural frequency, the phase ofsample zero from ADC 428 will not be zero. This leads to the use of themeasured marker phase as an error feedback term and a digital phaselocked loop to adapt the excitation frequency until the transmitterburst frequency precisely matches the resonant frequency of the marker.In this sense (when other factors are equal), marker locatorsimplemented according to embodiments of the present invention providesignal strength measurements at the maximum possible SNR for a givenantenna 406 and amplifier chain (including amplifier 426, filter 427,and ADC 428).

The same applies to the case when multiple marker types are stimulatedsimultaneously. As described above, a superposition of sinusoidalsignals is created at the transmitter output DAC 403, each component ofwhich has zero phase when referenced at ADC 428 input at time zero. Thusmultiple independent DPLLs could be implemented in detection DPLL 433 tosimultaneously track each type of marker detected in the environmentbased on the same simultaneous measurement. In practice, the resonantresponses of the various marker types generally overlap in frequency, soa more complex successive cancellation scheme can be implemented, as isfurther described below.

The signal processing steps necessary to implement the frequencyadaptation algorithm are now described. Although many of the algorithmsare described in this disclosure with references to block diagrams, thealgorithms can be implemented in software, hardware, or in someembodiments a combination of software and hardware. In some embodiments,algorithms described in this disclosure can be implemented on anintegrated circuit. The integrated circuit can include a microprocessorand memory to perform any or all of the functions described in thisdisclosure. Further, the integrated circuit may include dedicatedcircuitry for performing some or all of the functions described here.

A series of repetitive transmit pulses are captured at the receiver. Atypical pulse repetition rate for some embodiments of the presentinvention is between 500 and 1000 per second, with a duty cycle of25-50% (250 μs to 1 ms transmit burst time). By convention the parameterk is defined to represent the index of the transmit pulse since the lastfrequency change, while j defines the index of the ADC sample within anypulse k. By this definition j=0 at time zero of the k^(th) pulse(coincident with the first ADC sample from ADC 428). Then, according tosome embodiments of the present invention, the marker locator includes atime-domain averaging block 430 that enhances the coherent signalstrength of the marker(s) while significantly reducing the random noise.

FIG. 6 shows a block diagram of time averaging block 430 according tosome embodiments of the present invention. A time decay signal 600 and abuffer counter signal 607 can be received into time averaging block 430.Time decay signal 600, as can be seen from FIG. 4, is the sampled outputsignal from ADC 428. Buffer counter signal 607 is the pulse repetitionclock signal from marker transmit output 435 and represents the signalfor successive application of transmit output signal 422 to markerantenna 406.

In the embodiment shown in FIG. 6, time decay signal 600 is input todecay buffer 601. A running sum of the time decay signals after eachpulse can then be averaged in running sum block 602. Decay buffer 601buffers a selected number of samples corresponding to the length of themarker signal envelope decay after application of a pulse to markerantenna 406 and receipt of the return signal at marker antenna 406. Thevalues stored in decay buffer 601 are summed in running sum 602. Runningsum 602 performs a selected averaging method, where the averaging methodcan be either linear (unweighted sum over all pulses x(k)) orexponential (a first order recursive sum of the formy(k+1)=(1−A)x(k)+Ay(k), where A can be on the order of 0.99 in timeaveraging block 430). Latch output 606 latches latch 603 after receiptof N clock pulses, indicating that N transmit output pulses have beenapplied to marker antenna 406. After N pulses, the averaging sumcompiled in running sum 602 is latched in latch 603 and provided tobuffer 604 for serial output at average decay signal 605. As isapparent, but not shown in FIG. 6, timing is performed utilizing asystem clock that is much faster than the rate of pulses.

The average time N that triggers latch output 606 into latching latch603 can be defined as either the number of pulses that are linearlyaveraged together (linear average) or the number of pulses moved throughthe recursive sum (exponential average). In either case, after Ntransmitter pulses received, a normalized and averaged decay block isthe output from time averaging block 430. The length of the block insamples, i.e. the depth of buffers 601 and 604, is sufficient to capturethe entire decay of the marker(s) to the now lower noise floor, due tocoherent averaging. This depth can correspond to the entire time betweensuccessive transmitter pulses. In some embodiments, the linear averageis reset to zero after N pulses, which is usually at the time of afrequency update to the transmit block. In some embodiments, exponentialaveraging is not reset.

Some electronic markers may have memorized information (serial numbers,user data, position information) that is encoded into the marker decayresponse to the transmit burst. In one such example of a “smart marker”,the logic zero level can be represented by the absence of a markerresponse to a transmit burst and the logic one is represented by anormal response to the transmit burst. In other words, the smart markerchooses to blank its own response or not based on the information streamto be sent to the surface. The above ground marker locator detects thisbit stream by detecting the presence or lack thereof of a markerresponse per each transmit burst. It is still useful to enhance thecoherent signal strength of the smart marker by averaging, but thepresence or absence of the marker response utilize a more sophisticatedaveraging scheme. Since the marker locator at the surface has a prioriknowledge of the length of the data packets sent to it in the form of onor off marker responses, an array of average buffers could be allocatedto form one decay response average per bit. For example, as in FIG. 15,if the data packet consists of 3-bits, then the marker locator receiverwould allocate 3 time average buffers that allow capture of the entiresequence of three marker stimulus/responses 1507. Then a repeatedcommand of the smart marker to reply with the same binary data packetcould be averaged into an array of three buffers using the methoddescribed above. Each averaged buffer then will result in a higher SNRper data bit than would be achievable without time averaging of themarker decay response. In this example, decay responses 1502, 1504, and1506 (in response to stimuli 1501, 1503, and 1505) are decoded afteraveraging to represent the binary sequence “1-0-1”.

The averaged decay output signal 605 is then input to marker searchblock 431, a block diagram of an embodiment of which is shown in detailin FIG. 7. In FIG. 7, the averaged decay output signal 605 is depictedas decay signal 700. Marker search block 431 is enabled only if theenable bit “DPLL not locked” 701 is true. This enable signal isgenerated in Detection DPLL 433 and is further discussed below. Markersearch block 431 determines whether a marker or set of markers may bepresent in the environment, and forms a marker structure 708 (orpossibly an array of structures when multiple marker types may exist)for use in the remainder of the detection algorithm. The key informationcaptured in marker structure 708 is the marker frequency, phase at thatfrequency, and amplitude. When the DPLL is locked, structure 708 isalready initialized, and block 431 need not execute.

Marker search block 431 receives the time average signal from timeaverage block 431 as decay signal 700. Decay signal 700 is received infast-Fourier transform 702 that performs a linear Fourier transformfunction on decay signal 700. The results of the Fourier transform areinput to a determine active marker block 707 and may be output as partof marker structure signal 708. Initially, the marker frequency, phase,and amplitude can be derived from a thresholding operation on the outputsignal from linear FFT 702 of the time averaged signal 700. A goodindicator of the presence of one or more marker types results from theextraction of peaks from the spectrum that are close to the nominalmarker center frequencies. Amplitude and phase information are takenfrom the FFT result and written to the marker structure 706. Thecalibrated phase offset at that frequency is also carried along inmarker structure 706.

FIG. 8 shows an embodiment of detection DPLL 433. As shown in FIG. 4,detection DPLL 433 receives the average decay signal 605 from timeaverage 430 into decay signal 800 and the marker structure signal 708from marker searcher 431 into marker structure signal 708. A separatemarker analysis block is utilized for each active marker recognized. Inthe embodiment shown in FIG. 8, the marker structure signal 708 anddecay signal 800 are input to marker analysis block 801. Second andthird markers are analyzed in marker analysis blocks 802 and 803,respectively. In general, there can be any number of marker analysisblocks included in detection DPLL 433 so that any number of markers canbe simultaneously detected. For simple marker searches (when only onemarker type is present), blocks 802 and 803 are inactive since markerstructure 708 includes only one element.

A representative detection problem for a simple case is illustrated byFIG. 14A (Case 1, trace 1400), which shows the marker M1 transferfunction. FFT operation 702 leads to an initial determination that asingle marker type exists and that the estimated marker frequency is atfrequency 1405 on trace 1400. An error in the marker resonant frequencyestimate may occur because of the fixed frequency resolution of the FFT.For such a simple single marker case there is no real advantage to usingFFT 702 to predetermine the presence or absence of a marker. Markeranalysis block 801 will just track to the single marker resonantfrequency from any arbitrary starting frequency, even if frequency 1405is unknown. However, as will be shown below, FFT 702 is important wheresuccessive detection (i.e., detection of more than one marker) isutilized. Therefore, in some embodiments FFT 702 is included for theanalysis of simple cases as well.

FIG. 9 shows a block diagram of an embodiment of marker analysis block801, for example. Marker analysis blocks 802 and 803 (along with anyother marker analysis blocks that may be present) can have the same orsimilar structure as that depicted in FIG. 9. In the embodiment ofmarker analysis block 801, decay signal 900 (which is decay signal 800received in detection DPLL 433) and marker structure signal 708 is inputto digital phase-locked loop (DPLL) 901. The initial frequency estimate1405 (FIG. 14A) is taken from the frequency estimate provided in markerstructure signal 708.

A block diagram of an embodiment of DPLL 901 is shown in FIG. 10. Theembodiment of DPLL 901 illustrated in FIG. 10 is configured as a CostasLoop that relies on the phase error signal 1009 to act as the markerresonant frequency adaptation mechanism. The input decay sample 1000(which, in marker analysis block 801 corresponds to decay signal 800)and the initial frequency read from marker structure signal 708 areinput to numerically controlled oscillator (NCO) 1002. Because the inputaveraged decay signal 1000 is a block of samples, for example 500samples, the numerically controlled oscillator (NCO) sine and cosinegeneration is performed over all samples at the same frequency (i.e., nofrequency adaptation occurs inside of one decay signal block). The errorsignal 1009 is derived once per input block 1000 after quadraturemultiplication 1006, in-phase multiplication 1007, filtering operations1004 and 1005 that computes the mean value over all samples in theblock, and an inverse tangent operation 1008 to compute the phase angle.In the embodiment shown in FIG. 10, DPLL 901 generates one error signalper averaged decay input 1000, so that the next set of transmit pulseshas a shifted frequency according to the adaptation process. Once thefrequency has converged to substantially the natural frequency of themarker, the error term becomes effectively zero. At lock, the quadraturecomponent of the Costas Loop (Q) goes to zero and the inphase component(I) is taken as the marker signal strength 1010.

The actual frequency adaptation occurs in the NCO 1002, which results ina new current frequency 1011 for use in the transmitter outputgeneration later. FIG. 11 shows the structure of NCO 1002, which onlyruns if error signal 1009 is not close to zero, as determined by test1104. Before the lock condition, the NCO ramps the frequency accordingto the loop update equation:f(k+1)=f(k)+αe(k)  (Equation 2),where f(k) and e(k) represent the frequency and phase error of thecurrent decay block. Equation 2 is implemented by multipler 1105, summer1008, and feedback 1107. The parameter α is the feedback coefficient andis selected in accordance with the pulse repetition rate and, for manysystems, is about 0.00015.

The remainder of the first order NCO implements the equationsin(θ(k+1))=sin(2πfΔt+θ(k))  (Equation 3)for all the samples j residing in the input averaged decay signal. Thesine and cosine blocks 1110 and 1111 in FIG. 11 can be replaced withcorresponding lookup tables so that the real-time efficiency of NCO 1002can be improved. Likewise, multipliers 1103 and 1105 can be implementedusing fixed-point arithmetic processors for the same reason. Arithmeticissues are also important in the update of θ(k+1) in summer 1106 in FIG.11. If the phase angle θ(k) were allowed to increment over the entiredecay signal, the accuracy of NCO 1002 may suffer since a computer-basednumber system cannot represent unbounded numbers accurately. Thus, apractical implementation of NCO 1002 may include a modulo 2π circuit tocontrol the growth of θ in summer 1106.

Amplifier 1112 provides a gain of −1 applied to the output signal fromsine block 1110. Inverting the sine function creates negative feedbackof the phase error term and thereby drives the system to a lock state.Although digital phase locked loops generally operate on a continuousstream of data, there are more than enough samples in the averaged decaysignal 1000 to create a new estimate of phase error 1009.

When only a single marker type is present, there is no need forsuccessive detection so block 904 (FIG. 9) can be skipped and outputsignals 908 and 909 are updated with the latest estimates of markersignal strength 1010 and marker frequency 1011 from marker DPLL 901.

To close the loop with the new frequency estimate, the next transmitoutput vector is created in block 804 (FIG. 8). This repeats theprocedure described above with the additional option of implementingpower control 810 on the individual marker excitation frequencies. Thepower control option will be further discussed below in the discussionof successive detection, but this is also where the user interface canspecify the set of markers that are active 809.

The above discussion has illustrated the process of frequency adaptationfor a simple single-marker scenario. Over repeated loops, the adaptationmoves the current frequency along the transfer function curve 1406 (FIG.14A), until the phase error is zero at which the current frequency isthe marker resonant frequency. Even though the frequency updates onlyafter a new averaged decay signal 605 is available, the pulse repetitionrate (which, in some embodiments, is about 1000/sec) is high enough thatthe update occurs on the order of 10-20 times per second. With such areduction in noise floor (100 averages represents a factor of 10improvement in the SNR), as well as an appropriate choice for thefeedback parameter α 1003, a rapid lock time to a newly detected markeris achievable. At lock, the marker is tracked until the phase error 1009indicates that contact with that marker has been lost—for example whenthe operator has moved the marker locator out of range.

Successive cancellation detection can be utilized to positively decouplemarker responses when the “neighbor detection” and the “near-far”problems are present. Prior art methods attempt to surmount theambiguities by sequencing the marker locator transmitter pulse betweenthe various marker frequencies to be searched. An example of the“neighbor detection” problem is illustrated in FIG. 14B, Case 2. In FIG.14B, Case 2 is a situation in which three overlapping marker responsesexist, so that at frequencies F1 and F3 marker M1 and M3 responses 1401and 1403 are biased high as shown by combined response 1404, due to thepresence of a marker M2 at frequency F2. Similarly response 1402 isbiased by the resonant responses of markers M1 and M3 at frequencies F1and F3, respectively, to stimulus frequencies F1 and F3 that decay atnatural frequency F2 due to the frequency response of marker M2. Thesebiases will cause errors in the marker depth estimates. Even if thelocator sequences three successive transmitter pulses having separatesingle frequency stimuli—F1, F2, and F3—the same bias will result in thesignal strength measurements. It is important to note that during thedecay after the interruption of the transmitter burst, the response ofthe marker M1 at frequency F1 to stimulus F2 is a decaying time envelopat frequency F1 and initial amplitude equal to the M1 marker transferfunction at frequency F2.

Successive cancellation is a mechanism to remove the ambiguity ofwhether adjacent markers are present which can either bias the magnituderesponse of a targeted marker, or obscure its detection altogether. Thefollowing description illustrates an embodiment of a method ofcancellation according to the present invention. Beginning withsimultaneous transmission of a superposition of three stimulusfrequencies (F1, F2, F3, all equal in amplitude), and the subsequentacquisition of the averaged decay block 605 (as before), successivedetection begins with the marker search process 431, which isillustrated in FIG. 7. In the case illustrated in FIG. 14B, the resultof FFT 702 can be utilized to estimate the relative magnitudes of themarker signal at each of the three nominal marker frequencies. Ifnominal marker frequencies adjacent to the targeted marker aredetermined to have significant responses that stand out from the noise,marker structure signal 708 is initialized with each suspected markertype, including the initial frequency and phase that result from thelinear FPT analysis of FFT 702. The FFT signal from FFT 702 is alsouseful to determine the difference in the relative amplitudes, which forCase 2 (FIG. 14B) will show three marker types, or at least a broadregion of reflected energy that most likely results from a conditionthat multiple marker types are present within the frequency region fromF1 to F3.

Detection DPLL 433, which is further illustrated in FIGS. 8, 9, 10, and11 implements full successive detection mode, in that the output signalsof the dominant marker detection block 801 is a modified decay envelope910 generated by successive detector block 905, from which all responsesattributable to the dominant marker have been removed from the signal.For Case 2, Marker M2 is anointed the dominant marker since themagnitude response is highest at frequency M2. Once the error convergesto near zero for DPLL 901 (FIG. 9) relay 906 forces the “not locked”output 907 low, indicating to the rest of the system that a lock hasbeen established on the dominant marker.

The transition in the system from “not locked” to “locked” state canalso trigger the single degree of freedom (SDOF) curvefit operationperformed in SDOF curvefit block 904. The system parameters of a SDOFsystem (as represented by the marker LC resonant circuit), can bedetermined from three measurable quantities: the amplitude, the resonantfrequency f, and the 3 dB bandwidth of the response around the resonance(Δj). From these quantities can be determined the system equation, orequivalently the pole zero model of the system. The pertinent parametersfor the present purpose of successive detection are the amplitude,frequency, and decay time constant τ. For a SDOF system, a directequivalency between τ and Δf exists: $\begin{matrix}{\tau = \frac{1}{{\pi \cdot \Delta}\quad f}} & \left( {{Equation}\quad 4} \right)\end{matrix}$

To estimate Δf, two buffers 902 and 903 are accessed which contain thepast history of amplitude and frequency values that preceded the lockcondition. Each of those values represents the state of DPLL 901 overthe same interval (say, for example 100 averages), so a smooth estimateof the marker response is analyzed to determine the 3 dB bandwidth. Theonly condition necessary for the Δf calculation is that the oldeststarting frequency in the buffer is sufficiently away from the resonantfrequency that the amplitude values in the buffer differ by at least 3dB. Even when this condition is not met, the DSP controller 420 candecide to hold off the lock condition for some moments and shift thetransmit frequency in order to capture one more amplitude average so asto accomplish the SDOF curvefit operation (Equation 4).

The output signals from SDOF curvefit 904 include a set of parametersfrom which the time decay at frequency M2 can be synthesized insuccessive detection operation 905 (FIG. 12). Since averaged decay 900is a set of time domain samples, synthesis operation 1206 creates a timedomain block at the same sample-rate as decay 900. The starting phase ofthe synthesized block is zero (such as is the case for a frequency lockcondition of DPLL 901), and the starting amplitude is as specified byamplitude signal 1201. The frequency of the synthesized block is fixedfor the duration, and is substantially the resonant frequency of themarker at frequency F1. After subtraction 1207 the resulting modifieddecay 1208 is effectively cleaned of the dominant marker response atfrequency M2. The input signals amplitude 1201, frequency 1202, anddamping 1203 are received from SDOF curvefit 904 while marker structure1204 and decay 900 are received as input decay 900 and marker structure708 to marker analyzer 801.

As might be noted by inspection of the marker magnitude responses inFIG. 14 (Case 2), it may be difficult to achieve a lock condition in thefirst DPLL 901. This can happen if the overlapped responses are verysimilar in amplitude, such that no clear dominant marker response can bedetermined. Another potential problem is that DPLL 901 does notconverge, but that a marker center frequency error exists due to thefact that the zero of the aggregate phase of the time decay response isbiased due to the presence of more than one marker (almost equal inreturn energy).

Therefore, in some embodiments power control of the individual markerscan be accomplished. By adjusting the transmitted signal amplitudebetween the targeted marker type and the adjacent markers, a dominantmarker frequency can be more clearly identified. FIG. 14C, Case 3,represents a scenario that could result for the same physical placementof three markers (relative to the marker locator) as for Case 2. But nowthe amplitude of the marker M2 stimulus frequency F2 has been increasedrelative to the other two frequencies F1 and F3. In some embodiments,the F1 and F3 amplitudes can be decreased somewhat, to keep the overalltransmitter power output constant. In either case, through power controlthe probability of accurate detection of marker M2 has been enhanced atthe expense of identifying markers M1 and M3. Furthermore the frequencyerror due to any aggregate phase bias in the decay response has beenreduced. And even though the relative strength of markers M1 and M3 havebeen reduced, after successive cancellation of marker M2 the modifieddecay signal 910 has been more effectively cleaned of M2 signalcomponents. Thus the probability of detection of the M1 and M3 markershas improved, in the sense that the M1 to M2 signal-to-interferencenoise ratio (SNR) has increased (and likewise for the M3 to M2 SNR).Thus, power control at the transmitter adds an effective mechanism forthe marker locator to compose a signal by superposition that canstimulate an environment composed of multiple electronic markers(possibly having closely spaced frequencies) and can lead to increasedprobabilities of detection.

After the detection of the second marker M1 marker analysis block 802(FIG. 8), and possibly the third marker M3 in marker analysis 803, aftersuccessive cancellation of the responses from both markers M1 and M2,power control algorithm 810 can adjust the amplitude of the nexttransmitter output 804 based on the factors described above. Thesuperposition (addition of each signal) also occurs in transmitteroutput 804, as before using a reversed time method because of thenecessity of creating a combined output block wherein each and everysignal component results in a phase zero referred to ADC 430. To controlthe growth in amplitude of the last few samples before the zero phasereference point, the superposition method can optionally add eachsuccessive marker stimulus signal with either a 0° or 180° offset.Later, in the successive detection process at the output of the DPLL901, the error 1009 is negated for those signals that were added with a180° offset. Signal 808 represents the final transmit signal, which ispassed to the transmitter buffer through switch 434 (FIG. 4). Processor420 controls switch 434 to select calibration data 432, or the updatedsuperposition of amplitudes and frequencies, based on the measurementstate. FIG. 13 shows a diagram of the final stage before writing thebuffer sample by sample to the output DAC 403. A time reversal 1304(buffer direction swap) of the received transmit output signal 1301 andbuffer 1305 is executed so that the phase zero point is appropriatelyaligned, and the pulse repetition clock is aligned at the buffer edges.A time reversed transmit output signal 1307 and a pulse repetition clock1306, which is generated by transmit pulse generator 1302, are output bytransmit output block 435.

The embodiments described herein are examples only of the invention.Other embodiments of the invention that are within the scope and spiritof this disclosure will be apparent to those skilled in the art fromconsideration of the specification and practice of the inventiondisclosed herein. It is intended that the specification and examples beconsidered as exemplary only and not limiting. The scope of theinvention, therefore, is limited only by the following claims.

1. A marker locator, comprising: at least one transmitter channelcoupled to provide a transmitter signal to an electromagnetic fieldgenerator, wherein the transmitter signal is phase shifted tosubstantially zero and is capable of exciting multiple types of markers;at least one receiver channel coupled to receive time decay signals froman electromagnetic field detector; and a digital processor coupled toprovide digitized transmitter output signals to the at least onetransmitter channel and to receive digitized time decay signals from theat least one receiver channel, wherein the digital processor receivesthe digitized time decay signals from the at least one receiver channelin the time interval between pulses of application of digitizedtransmitter output signals to the at least one transmitter channel, theaverages the digitized time decay signals received from the at least onereceiver channel, adapts the frequency of the averaged time decaysignals, wherein the adapted frequency and the averaged time decaysignals can assist in distinguishing between the multiple types ofmarkers.
 2. The locator of claim 1, wherein a pulse includes a firstperiod where transmission of the transmitter signal from theelectromagnetic field generator occurs and a second period where thetime decay signal can be received.
 3. The locator of claim 1, whereinthe at least one transmitter channel includes a digital-to-analogconverter coupled to receive a digitized transmitter output signal fromthe digital processor, a filter coupled to receive signals from thedigital-to-analog converter, and a driver coupled to receive signalsfrom the filter and provide the transmitter signal to theelectromagnetic field generator.
 4. The locator of claim 1, wherein theelectromagnetic field generator and the electromagnetic field detectorare an antenna, and wherein the at least one transmitter channelincludes a transmit switch and the at least one receiver channelincludes a receive switch, the transmit switch and the receive switchbeing controlled by the digital processor such that the at least onereceiver channel does not receive the time decay signals while the atleast one transmitter channel is coupled to the antenna and the at leastone transmitter channel does not transmit the transmitter signal whenthe at least one receiver channel is coupled to the antenna.
 5. Thelocator of claim 1, wherein the at least one receiver channel includes afilter coupled to an analog-to-digital converter, the filter coupled tothe electromagnetic field detector to receive the time decay signals andthe analog-to-digital converter coupled to the digital processor toprovide digitized time decay signals.
 6. The locator of claim 1, whereinthe digital processor includes a time average that receives thedigitized time decay signals from the at least one receiver channel, thedigitized time decay signals indicating a signal decay as a function oftime following a pulse of the transmitter signal to the at least onetransmitter channel, the time average averaging signals over apredetermined number of time intervals between pulses to generate anaverage decay signal; a marker search that determines a marker structurefrom the average decay signal generated by the time average, the markerstructure including parameters characterizing at least one marker thatreflects signals detected by the electromagnetic field detector; adetection digital phase locked loop that receives the marker structure,separates a contribution from each of the at least one marker,determines marker parameters for each of the at least one marker, andgenerates a transmit array; and a marker transmit output that generatesthe digitized transmitter output signal of the pulse based on thetransmit array and a pulse repetition clock.
 7. The locator of claim 6,wherein the time average utilizes a linear averaging.
 8. The locator ofclaim 6, wherein the time average utilizes an exponential averaging. 9.The locator of claim 6, wherein the time average comprises: a decaybuffer that stores the digitized time decay signals received from the atleast one receiver channel; a summer that adds the signals stored in thedecay buffer so that an average is formed over a preselected number ofpulses; a latch that latches that latches signals from the summer intoan output buffer to provide the average decay signal; and a latch outputthat counts the number of pulses and latches the latch after receipt ofthe preselected number of pulses.
 10. The locator of claim 6, whereinthe time average comprises a plurality of decay buffers to storedigitized time decay signals corresponding to a plurality of bits ofdata from at least one smart marker; a plurality of summers that add thesignals stored in each of the plurality of decay buffers so that anaverage is formed over a preselected number of pulses; a plurality oflatch outputs that latch signals from the summer into a plurality ofoutput buffers to provide an average decay signal corresponding witheach of the plurality of bits; and a plurality of latch outputs thatcounts the number of pulses and latches a corresponding one of theplurality of latch outputs after receipt of the preselected number ofpulses.
 11. The locator of claim 6, wherein the marker search comprises:a fast fourier transform that receives the average decay signal from thetime average and provides a fast fourier transform signal of the averagedecay signal; a marker determiner coupled to receive the fast fouriertransform signal and generate a marker structure corresponding toparameters indicating each of at least one identified marker
 12. Thelocator of claim 11, wherein the marker structure can include the markerfrequency, the marker amplitude, and the marker phase of the at leastone identified marker.
 13. The locator of claim 6, wherein the detectiondigital phase locked loop is coupled to receive the average decay signaland the marker structure, the digital phase locked loop comprising: atleast one marker analyzer coupled to generate a marker strength andmarker frequency for a marker and further to remove the effects of thatmarker from the decay signal, each of the marker analyzers being coupledto receive a decay signal from a previously coupled digital phase lockedloop with a first digital phase locked loop coupled to receive theaveraged decay signal; and a transmit output generator, the transmitoutput generator providing a transmit array for determining a digitaltransmit signal.
 14. The locator of claim 13, wherein each of the markeranalyzers includes: a digital phase locked loop coupled to receive adecay signal and the marker structure, the digital phase locked loopconverging on a current frequency and providing a signal strength, andan error signal corresponding to one marker indicated in the markerstructure; a SDOF curvefit coupled to receive the current frequency andsignal strength and generate a marker strength and marker frequency; anda successive detector generation coupled to receive signals from theSDOF curvefit, generate a marker response, and subtract the markerresponse from the decay signal to generate a new decay signal forprocessing by another one of the at least one marker analyzers.
 15. Alocator of claim 13, wherein the marker transmit outputs receives thetransmit array and provides a time reversal to generate the digitaltransmit signal.
 16. The locator of claim 15, wherein the digitaltransmit signal includes signals at frequencies centered around resonantfrequencies of each of a plurality of markers.
 17. The locator of claim16, wherein the transmit array is arranged to have 0 phase shift foreach of the resonant frequencies as measured at an input to the timeaverage. 18.-24. (canceled)
 25. A marker locator, comprising: means forgenerating pulses of a transmit signal from a frequency, field strength,and phase from one or more markers, wherein these pulses can excitemultiple types of markers and each type of marker represents a differenttype of utility line; means for receiving signals from one or moremarkers between applications of the transmit signal; means for averagingthe received signals to form an average decay signal; means fordetermining the frequency, field strength, and phase of one or moremarkers from the average decay signal, wherein the determined frequencythe determined field strength, and the determined phase can assist indistinguishing between multiple types of markers.
 26. The marker locatorof claim 25, wherein the means for generating pulses includes means fordetermining a marker transmit signal for at least one active marker; andmeans for generating a digital transmit signal from the marker transmitsignal for each of the at least one active marker.
 27. The markerlocator of claim 26, wherein the means for determining the markertransmit signal includes a means for adjusting a power of the markertransmit signal for each of the at least one active marker.